Active resistance summer for a transformer hybrid

ABSTRACT

A transmit canceller comprises an operational amplifier having a first polarity input terminal, a second polarity input terminal, and an output terminal. A feedback element communicates with the second polarity input terminal and the output terminal. A first input resistor communicates with the second polarity input terminal and the measured signal input. A second input resistor communicates with the second polarity input terminal and the replica signal input. A predetermined voltage source communicates with the first polarity input terminal of the operational amplifier. The received signal is an output at the output terminal of the operational amplifier.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.10/786,010 filed on Feb. 26, 2004, which is a Continuation of U.S.patent application Ser. No. 09/629,092 filed Jul. 31, 2000 now U.S. Pat.No. 6,775,529. The disclosures of the above applications areincorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to transmitting and receivingelectrical signals through communication channels, such as a gigabitchannel. In particular, the present invention relates to a transmitcanceller that removes transmit signals from receive signals in suchcommunication channels.

2. Background and Related Art

A gigabit channel is a communications channel with a total datathroughput of one gigabit per second. A gigabit channel typicallyincludes four (4) unshielded twisted pairs (hereinafter “UTP”) of cables(e.g., category 5 cables) to achieve this data rate. IEEE Standard802.3ab, herein incorporated by reference, specifies the physical layerparameters for a 1000BASE-T channel (e.g., a gigabit channel).

As will be appreciated by those skilled in the art, a UTP becomes atransmission line when transmitting high frequency signals. Atransmission line can be modeled as a network of inductors, capacitorsand resistors, as shown in FIG. 1. With reference to FIG. 1, G isnormally zero and R(T) is complex due to skin effect. R(T) can bedefined by:R(ω)=k _(R)(1+j)√{square root over (ω)}  (1)where k_(R) is a function of the conductor diameter, permeability, andconductivity. The characteristic impedance of the line is defined by:

$\begin{matrix}{{Z_{0} = \sqrt{\frac{{R(\omega)} + {{j\omega}\; L}}{G + {{j\omega}\; C}}}},} & (2)\end{matrix}$

and at high frequencies, Z₀ becomes approximately √{square root over(L/C)} or approximately 100 ohms in a typical configuration. Whenproperly terminated, a UTP of length d has a transfer function H that isa function of both length (d) and frequency (ω):H(d,ω)=e ^(dγ(ω)),  (3)whereγω=√{square root over ((R(ω)+jωL)(G+jωC))}{square root over((R(ω)+jωL)(G+jωC))},  (4)

and substituting Equations 1 and 4 into Equation 3, and simplifying,approximately yields:

$\begin{matrix}{{H\left( {d,\omega} \right)} \approx {\left\{ {d\left\lbrack {{\frac{k_{R}}{2}\sqrt{\frac{\omega\; L}{C}}} + {j\left( {{\omega\sqrt{L\; C}} + {\frac{k_{R}}{2}\sqrt{\frac{\omega\; L}{C}}}} \right)}} \right\rbrack} \right\}.}} & (5)\end{matrix}$Equation 5 shows that attenuation and delay are a function of the cablelength d.

A transmission path for a UTP typically includes a twisted pair ofcables that are coupled to transformers at both a near and far end, asshown in FIG. 2. A transceiver at each end of the transmission pathtransmits and receives via the same twisted pair. A cable typicallyincludes two patch cords totaling less than 10 m, and a main section of100 m or even longer. The transmitters shown in FIG. 2 are modeled ascurrent sources. The near end current source supplies a current I_(tx).The near end transmit voltage (e.g., I_(tx)R_(tx)) is detected andmeasured across resistor R_(tx). A receive signal V_(rcv) (e.g., asignal transmitted from the far-end transceiver) is also detected andmeasured across resistor R_(tx). Hence, V_(tx) includes both transmit(I_(tx)R_(tx)) and receive (V_(rcv)) signals. Accordingly, the signalV_(rcv) (e.g., the signal from Transceiver B) received at Transceiver Acan be obtained by taking the difference between the transmit voltageand the measured voltage V_(tx), as follows:i.V _(rcv) =V _(tx) −I _(tx) R _(tx).  (6)

Conventional solutions for removing transmit signals from receivesignals often employ known transconductor (“Gm”) summing stages or othercurrent based methods. As will be appreciated, these methods oftenintroduce signal distortion into the receive signal. Also, sometransconductors have a limited signal dynamic range. Accordingly,conventional methods are often inadequate for applications requiringsignal recovery. Additionally, known summing circuits, such as weightedsummers using operational amplifiers, have not heretofore been modifiedto accommodate the intricacies associated with canceling transmitsignals or regulating baseline wander (described below). A knownweighted summer is discussed in Chapter 2 of “Microelectronic Circuits,Third Edition,” by A. S. Sedra and K. C. Smith, 1991, incorporatedherein by reference.

As will be appreciated by those skilled in the art, the receive signalV_(rcv) typically contains additional components, due to baselinewander, echoes and crosstalk, for example.

Baseline wander is preferably corrected for when transmitting andreceiving signals over transmission lines. Removing DC components from areceive signal using transformer coupling can cause baseline wander. Aswill be appreciated by those skilled in the art, baseline wanderrepresents a deviation from an initial DC potential of a signal.

“Echoes” typically represent a residual transmit signal caused byreflections that appear in the receive signal. Echoes can cause undueinterference depending on the size of the reflection.

Capacitive coupling between the channels, as shown in FIG. 3, causescrosstalk. Four channels TX1–TX4 are shown in FIG. 3. The capacitivecoupling between TX1 and each of TX2, TX3 and TX4 are modeled bycapacitors C₁₋₂, C₁₋₃, C₁₋₄, respectively. The capacitive coupling formsa high-pass filter between channels and therefore crosstalk containsmostly high frequency components. As will be appreciated by thoseskilled in the art, normally only the near-end crosstalk (NEXT) needs tobe considered, since crosstalk is usually small and the transmissionline provides further attenuation of the far-end crosstalk (FEXT).

Accordingly, there are many signal-to-noise problems to be solved in theart. Hence, an efficient transmission canceller is needed to remove atransmit signal from a receive signal without introducing excess signaldistortion. An electrical circuit is also needed to subtract a transmitsignal from a receive signal. There is a further need of an electricalcircuit to correct baseline wander.

SUMMARY OF THE INVENTION

The present invention relates to a transmit signal canceller for use ina transformer hybrid. Such a hybrid includes a junction for transmittingand receiving signals. In the present invention, an active resistivesummer can be used to cancel a transmit signal from a receive signal.

According to the invention, an electrical circuit in a communicationschannel is provided. The electrical circuit includes an active resistivesummer having: (i) an input for a composite signal, the composite signalincluding a transmission signal component and a receive signalcomponent, (ii) an input for a replica transmission signal, and (iii) anoutput for a receive signal which includes the composite signal minusthe replica signal.

According to an another aspect of the present invention, a transmitsignal canceller in a communication channel is provided. The channelincludes a first transceiver for transmitting and receiving signals anda replica transmitter for generating a replica transmission signalinput. A composite signal at a rear end includes a transmission signalof the first transceiver and a received signal of a second transceiver.The transmit canceller includes: (i) an operational amplifier having apositive input terminal, a negative input terminal, and an outputterminal; (ii) a feedback element in communication with the negativeinput terminal and the output terminal; (iii) a first input resistor incommunication with the negative input terminal and the measured signalinput; (iv) a second input resistor in communication with the negativeinput terminal and the replica signal input; and (v) a predeterminedvoltage source in communication with the positive terminal of theoperational amplifier. The receive signal is an output at the outputterminal of the operational amplifier.

According to still another aspect of the present invention, acommunication system including a first transmission channel with a firstend and a second end is provided. The first end couples to a firsttransformer and the second end couples to a second transformer. A firsttransceiver transmits and receives signals via the first transformer anda second transceiver transmits and receives signals via the secondtransformer. A first signal is supplied at the near end. The firstsignal includes a transmission signal component of the first transceiverand a receive signal component of the second transceiver. Thecommunications system includes: (i) a replica transmitter that generatesa replica of the transmission signal component of the first transceiver;(ii) a filter to filter the replica signal; (iii) an active resistivesummer receiving the first signal, and the filtered replica signal asinputs to reduce the transmission signal component at an output of theactive resistive summer.

According to still another aspect of the present invention, a method ofcorrecting baseline wander in a receive signal in a communicationschannel having a near and far end is provided. The channel includes afirst transceiver at the near end and a second transceiver at the farend, each to transmit and receive signals. The method includes the stepsof: (i) providing a composite signal, the composite signal including atransmission signal of the first transceiver and a receive signal of thesecond transceiver; (ii) generating a replica of the transmissionsignal; (iii) subtracting the replica signal from the composite signalthrough an active resistive summer; and (iv) providing a baselinecorrection current into the active resistive summer.

These and other objects, features, and advantages of the presentinvention will be apparent from the following description of thepreferred embodiments of the present invention.

BRIEF DESCRIPTION OF THE DRAWINGS

The details of the present invention will be more readily understoodfrom a detailed description of the preferred embodiments taken inconjunction with the following figures.

FIG. 1 is a circuit diagram illustrating a transmission line model.

FIG. 2 is a circuit diagram illustrating a transmission path across atwisted pair of cables, the cables being coupled to transformers at eachend.

FIG. 3 is a diagram-illustrating crosswalk between channels in a gigabitchannel.

FIG. 4 is a block diagram illustrating a system overview of acommunications channel.

FIG. 5 is a circuit diagram illustrating a transmitter.

FIG. 6 is a graph illustrating a transmit signal.

FIG. 7 is a graph illustrating a composite signal with echoes.

FIG. 8 is a circuit diagram illustrating a replica transmitter.

FIG. 9 is a graph illustrating a receive signal.

FIG. 10 is block diagram illustrating a low-pass filter.

FIG. 11 is a circuit diagram illustrating an active resistive summer.

FIG. 12 is a circuit diagram illustrating an error detection circuit.

FIG. 13 is a circuit diagram illustrating a low-pass filter.

FIG. 14 is a circuit diagram illustrating a conventional voltagecontrolled current source.

DETAILED DESCRIPTION OF THE PRESENTLY PREFERRED EMBODIMENTS

The preferred embodiments will be described with respect to a gigabitchannel, as used, for example, in an Ethernet network; and to electricalcircuits associated with separating transmit and receive signals in sucha gigabit channel. The preferred embodiments will also be described withrespect to baseline wander correction in such a gigabit channel.However, as will be appreciated by those skilled in the art, the presentinvention is also applicable to other transmission channels, and toother electrical circuits having applications requiring cancellation oftransmit signals, for example.

FIG. 4 is a block diagram illustrating principle components for one ofthe four channels in a preferred gigabit channel configuration for usein an Ethernet network. As illustrated in FIG. 4, a vertical dashed linedivides analog and digital processing components. The analog componentspreferably include a transmitter (“XMTR”) 1, replica transmitter(“Replica XMTR”) 2, transmit canceller 3, baseline correction module 4,low pass filter (“LPF”) 5, analog-to-digital converter (“ADC”) 6, andphase-lock loop (“PLL”) 7. A known PLL can be used with the presentinvention.

Digital processing components preferably include a transmitter encoder10, echo module 11, NEXT cancellers 12–14 to assist in removing echoes,synchronization module 15, FIR (Finite Impulse Response) equalizer 16and a DFE (Decision Feedback Equalizer) 17 to equalize a receive signal,and a Viterbi module 18. The digital processing components also includebaseline correction modules 19 and 20 to correct residual baselinewander. A timing recovery module 21, an error correction detector 22(described in further detail below), and summing junction 23 are alsoshown. The individual digital components designated by blocks in FIG. 3are all well known in the communication arts, and their specificconstruction and operation are not critical to the operation or bestmode for carrying out the present invention.

The analog “front-end” components shown in FIG. 4 will now be describedin even further detail. The front-end analog components are preferablydesigned and constructed via customized integrated circuits. However, aswill be appreciated by those skilled in the art, the inventive circuitsand corresponding configuration could also be realized using discretecomponents as well.

As illustrated in FIG. 5, transmitter 1 preferably includes acurrent-source I_(tx) that generates a transmit signal over a resistorR_(tx). An appropriated value for resistor R_(tx) can be selected tomatch the line impedance, for example. In one preferred embodiment, aresistor center tap is set to 2.5 volts so the transmitter 1 effectivelysees a differential impedance of 25 ohms. Preferred performancespecifications for the transmitter 1 are further detailed in Table 1,below.

An impulse transmit signal can be generated from a unit square pulse of1T width filtered by a one-pole, low-pass filter (not shown) with acutoff frequency between 85 MHz and 125 MHz. Slew-rate control can alsobe used to limit the rise and fall times and thus reduce the highfrequency components of a transmit signal. Of course, any transmitsignal preferably fits into the transmit template provided by the IEEE802.3ab Standard. An ideal transmit pulse is shown in FIG. 6.

A measured voltage V_(tx) across R_(tx) (FIG. 5) is shown in FIG. 7. Themeasured signal V_(tx) contains interference caused by line reflections(e.g., echoes). The reflections are caused by impedance discontinuitydue to impedance mismatch between different cables. For example, a largereflection pulse at 60 ns as shown in FIG. 7 corresponds to a reflectionfrom the impedance discontinuity at an adapter connecting a 5 m patchcord to a 100 m cable. The magnitude of the echoes can be significantwhen compared to the magnitude of the receive signal at a long linelength, and therefore, echo cancellation, as provided by the NEXTcancellers 12–14 shown in FIG. 4, is employed.

A receive signal V_(rcv) (e.g., a signal received from a far-endtransceiver) is also measured across resistor R_(tx), as shown in FIG.5. Accordingly, the near end transmit signal (I_(tx)R_(tx)) ispreferably canceled or reduced from the composite signal V_(tx) in orderto effectively recover the far-end received signal V_(rcv). This type ofactive cancellation can be accomplished with a replica transmit signalV_(txr). Accordingly, a replica transmitter 2 (to be described below) isprovided to generate a signal V_(txr) to be subtracted from the measuredsignal V_(tx), thus, effectively reducing the transmit signal(I_(tx)R_(tx)).

A receive signal x(t) transmitted with pulse amplitude modulation(“PAM”) is define by:

$\begin{matrix}{{{x(t)} = {\sum\limits_{n = 1}^{\infty}\;{a_{n}{p\left( {t - {n\; T}} \right)}}}},} & (7)\end{matrix}$

where a_(n) is the transmit symbols and p(t) is the channel pulsederived by convoluting an impulse transmit pulse with a channel responsedefined by Equation 5. The receive signal for a 100 m cable is heavilyattenuated by the transmission line and the pulse width is dispersed, asshown in FIG. 9. A 100 m UTP delays the signal by about 550 ns. Signalequalization preferably uses high frequency boosting via the FIR 16 toremove precursor intersymbol interference (“ISI”) and to insert a zerocrossing for timing recovery 21. The DFE 17 is used to remove postcursorISI.

The receive signal's elongated tail results from transformer coupling(e.g., a high-pass filter) with a time constant (e.g., L/R) typically onthe order of micro-seconds. Since the receive signal contains little orno average DC energy, the negative tail has the same amount of energy asthe positive pulse. In this regard, the signal's area integral is zero.In a typical example, a tail can last over 10 μs with a magnitude of nomore than 0.5 mV. The long tail causes any DC bias to drift back towardzero, which can lead to baseline wander. As will be appreciated, thisresponse time is too long to be practically removed by a digitalequalizer, but the response is slow enough to be cancelled using a slowintegrator, for example. The baseline wander canceller 4 is preferablydecision directed to minimize the error defined by the differencebetween the equalized value and it's sliced value, as discussed below.

As illustrated in FIG. 8, the replica transmitter 2 includes a currentsource I_(txr). I_(txr) is coupled to a voltage V through resistors R,as shown in FIG. 8. In a preferred embodiment, R is 100 ohms and V isabout 2.5 volts. The replica signal V_(txr) is preferably filteredthrough a known low-pass filter to obtain a low-pass replica signal(“V_(txrl)”), as shown in FIG. 10. Replica signal V_(txr) can also beinverted in a known manner to produce −V_(txr). The preferredperformance specifications for the transmitter 1 and replica transmitter2 are shown in Table 1.

TABLE 1 Transmitter and Replica Performance Specifications ParametersSpecifications Transmit Current +/−40 mA Replica Transmit ¼ of transmitcurrent Current Number of levels 16 (not including 0) Number of sub- 8(sequentially delayed) units Transmit Profile [1 1 2 2 1 1], w/~1 nsdelay Replica Transmit [1 1 3 3], w/~1 ns delay Profile R_(tx) 100Ω

A transmit signal canceller 4 is illustrated in FIG. 11. The transmitcanceller 4 removes the transmission signal (I_(tx)R_(tx)) from themeasured (or detected) transmit V_(tx) signal. In particular, thetransmit canceller includes an active resistive summer that provides alarge input dynamic range and stable linearity characteristics, whileremoving (e.g., reducing or canceling) the unwanted transmit signalcomponent.

As illustrated in FIG. 11, the active summer includes an operationalamplifier (“op-amp”) with inverting feedback. The op-amp is preferablyconstructed using integrated circuits in a known manner. The summerreceives V_(txrl), V_(tx), −V_(txr), I_(cms), and I_(bl), as inputsignals. I_(bl) is a baseline wander control current, and I_(cms) is acommon-mode shift current, each as further discussed below.

As will be appreciated by those skilled in the art, a transformertypically has high-pass characteristics. Accordingly, replica signal−V_(txr) is combined (e.g., subtracted via the active resistive summer)with the low pass replica signal V_(txrl) to produce a high-pass replicasignal. As an alternative configuration, V_(txr) could be filteredthrough a known high-pass filter prior to the transmit canceller 3stage.

Returning to FIG. 11, receive signal V_(rcv) is determined from thefollowing relationships.

Let:

Vi=voltage for the op-amp's positive terminal;

V₁=V_(txrl);

V₂=V_(tx);

−V₃=−V_(txr);

i₄=I_(cms); and

i₅=I_(bl).

Then:

$\begin{matrix}\begin{matrix}{{{{(i){~~~}i_{1}} + i_{2} - i_{3} - i_{4} - i_{5}} = i_{0}};{and}} \\{{{\left( {i\; i} \right){~~~}\frac{V_{1} - {V\; i}}{R_{1}}} = i_{1}};{\frac{V_{2} - {V\; i}}{R_{1}} = i_{2}};{\frac{V\;{i--}V_{3}}{R_{1}} = i_{3}};{\frac{{{V\; i} - {V\; r\; c\; v}}\;}{R_{F}} = \left. {i_{0}.}\Rightarrow \right.}} \\{{\frac{V_{1} - {V\; i}}{R_{1}} + \frac{V_{2} - {V\; i}}{R_{1}} - \frac{{V\; i} - V_{3}}{R_{1}} - i_{4} - i_{5}} = \left. \frac{{{V\; i} - {V\; r\; c\; v}}\;}{R_{F}}\Rightarrow \right.} \\{{\frac{V_{1} + V_{2} - V_{3} - {3\; V\; i}}{R_{1}} - i_{4} - i_{5}} = \left. \frac{{{V\; i} - {V\; r\; c\; v}}\;}{R_{F}}\Rightarrow \right.} \\{{{\frac{R_{F}}{R_{1}}\left( {V_{1} + V_{2} - {V_{3}3\; V\; i}} \right)} - {i_{4} \cdot R_{F}} - {i_{5} \cdot R_{F}}} = \left. {{V\; i} - {V\; r\; c\; v}}\Rightarrow \right.} \\{{{\frac{R_{F}}{R_{1}}\left( {V_{1} + V_{2} - V_{3} - {3\; v\; i}} \right)} - {R_{F}i_{4}} - {R_{F}i_{5}} - {V\; i}} = -} \\{{V\; r\; c\; v\mspace{14mu} V_{r\; c\; v}} = {V_{i} - {\frac{R_{F}}{R_{1}}\left( {V_{1} + V_{2} - V_{3} - {3\; V\; i}} \right)} + {R_{F}\left( {i_{4} + i_{5}} \right)}}}\end{matrix} & (8)\end{matrix}$

Substituting the input signals for their placeholders yields thedefinition for V_(rcv), as follows:

$\begin{matrix}{V_{r\; c\; v} = {{V\; i} - {\frac{R_{F}}{R_{1}}\left( {{V\; t\; x\; r\; l} + {V\; t\; x} - {V\; t\; x\; r} - {3\; V\; i}} \right)} + {{R_{F}\left( {{I\; c\; m\; s} + {I\; b\; l}} \right)}.}}} & (9)\end{matrix}$

The gain is preferably set between 0.75 and 1 (e.g., R_(F)/R₁ equals0.75 to 1). For a small signal analysis, Vi can be set to zero (0).Also, as will be appreciated by those skilled in the art, in a fullydifferential circuit, Vi effectively drops out of the equations sinceV_(rcv)=V_(rcv) ⁽⁺⁾−V_(rcv) ⁽⁻⁾. As discussed, V_(txrl) and −V_(txr) arecombined through the active summer to provide a high-pass replica signal(“V_(txrh)”). The receive signal V_(rcv) can then be recovered as shownby Equation 9.

Preferred transmit canceller specifications are detailed in Table 2,below.

TABLE 2 Transmit Canceller Performance Specifications ParametersSpecifications Input Dynamic +/−2.5 V(diff.) for transmit signal RangeOutput Dynamic +/−1 V(diff.) Range Input impedance High, ~10 k. Outputimpedance Low Cutoff frequency Greater than 31.5 Mhz DC Gain 0.85-dependent on the LPF 5 and ADC 6 characteristics (FIG. 4) Power 25 mw,including LPF 5 (FIG. 4) R_(f) 8.5 KΩ; or 7.5 KΩ for increasedattenuation Vi 2.0 volts R₁ 10 KΩ

A known current mode circuit, e.g., a voltage controlled current source(VCCS) as shown in FIG. 14, with feedback preferably sets the summerinput current-mode voltage (V_(cm)). Of course, other known current modecircuits could be employed with the present invention. This current-modecircuit shifts the common-mode of both the transmit and replica transmitsignals. The input to the op amp (V_(aip), V_(ain)) is compared againstthe desired op amp output common-mode voltage (V_(d)):V _(d)=(V _(aip) −V _(cm))+(V _(ain) −V _(cm)).  (10)

Then, the common-mode shift current can be determined from:I _(cms) =V _(d) g _(m) +I ₀,  (11)where g_(m) is a transconductance and I_(o) is an offset current. Anappropriate transconductance and offset current can be selected bysetting V_(cm)=I_(cms)R_(F)=V_(d)g_(m)R_(F)+I₀R_(F), to ensure a propercommon-mode voltage seen by the op amp inputs. In this manner, thecommon mode shift current I_(cms) can be regulated to pull down thecommon mode voltage of the operational amplifier as needed.

Baseline wander current I_(bl) is also “summed” by the active resistivesummer, as shown in FIG. 11, to correct baseline wander. Approximatelyninety percent (90%) of all system baseline correction can be obtainedthrough the active summer. The remaining baseline residual can bedigitally corrected through an equalizer, for example. As will beappreciated, the FIG. 11 topology allows the current sources (I_(bl) andI_(cms)) to each have a fixed output voltage, thus, minimizing currentdeviation due to finite output resistance.

The baseline wander correction module 4 preferably corrects for baselinewander using a decision-directed method, such as a discrete integrator.The decision-directed method can be implemented with a known chargepump, where the pump sign (e.g., +1/−1) is determined digitally using anerror between the equalized baseline signal (y_(k)) and a slicedbaseline signal (y^_(k)), as shown in FIG. 12. As will be appreciated bythose skilled in the art, the expected error value (e.g., E[e_(k)]) isideally driven to zero. The charge pump is preferably pumped up or downbased on the error value. For example, a positive error implies that anegative value should be input into the charge pump. For a negativeerror, a positive value should be input into the charge pump. The chargepump preferably has at least two current settings to regulate I_(bl). Ofcourse, a charge pump with many current settings could be used to obtainfiner baseline correction control.

The preferred baseline wander correction performance specifications arefurther detailed in Table 3, below.

TABLE 3 Baseline Wander Correction Specification ParametersSpecifications Output Dynamic +/−100 uA (diff.), (+/−1 V/R₁, R₁ = 10 kΩ)Range Output impedance High Integration Factors 2 mV/T, 4 mV/TBandwidth >100 MHz

A second-order low-pass filter, as shown in FIG. 13, is cascaded afterthe summer to preferably flatten the frequency response out to about31.25 MHz (<1 dB). A minimum overall attenuation of 20 dB at 125 MHz isdesirable for the low pass filter. In a sampled system, some aliasingbeyond Nyquist frequency (or excess bandwidth) is acceptable, butminimum aliasing is allowed at the sampling frequency. The transmitteddata is preferably band-limited to the Nyquist rate.

Preferred performance characteristics of the low pass filter 5 arefurther detailed in Table 4, below.

TABLE 4 LPF Performance Specification Parameters Specifications InputDynamic +/−1 V(diff.) Range Output Dynamic +/−1 V(diff.) Range Inputimpedance High, ~10 k. Output impedance Low Cutoff frequency 50–60 Mhz.Q (2nd order) ~1 Input impedance High, ~10 k. Output impedance Low, <100DC gain 1

As an alternative arrangement, a third-order Sallen and Key low passfilter as disclosed in a co-pending application by the same inventor ofthis application, titled “CALIBRATION CIRCUIT,” filed concurrentlyherewith, and hereby incorporated by reference, could be used as filter5. Similarly, the calibration circuit disclosed therein could also beused to calibrate the low pass filter 5.

Analog-to-digital converters are well know in the art. As will beappreciated, the ADC 6 resolution is often determined by system digitalprocessing requirements. In a preferred embodiment, the Viterbi detector18 requires an effective 7-bit resolution. Residual baseline wander,echoes, and crosstalk increase the dynamic range by about 200–300 mV,which increases the required resolution. The reduction in dynamic rangedue to insertion loss for a 100 m cable is approximately 40%.Accordingly, an 8-bit resolution is preferred.

The preferred ADC performance specifications are further detailed inTable 5, below.

TABLE 5 ADC Performance Specification Parameters SpecificationsResolution 8-bits minimum. Sampling frequency 125 MS/s Source OutputLow, ~200–400Ω Impedance

Thus, a transmit canceller including an active resistive summer has beendescribed. Such an active resistive summer has not heretofore beendeveloped for applications such as canceling signals in gigabitchannels. Correcting baseline wander through such an active resistivesummer has also been described herein.

While the present invention has been described with respect to what ispresently considered to be the preferred embodiments, it will beunderstood that the invention is not limited to the disclosedembodiments. To the contrary, the invention covers various modificationsand equivalent arrangements included within the spirit and scope of theappended claims. The scope of the following claims is to be accorded thebroadest interpretation so as to encompass all such modifications andequivalent structures and functions.

For example, while preferred circuit configurations and component valueshave been described, it will be understood that modifications could bemade without deviating from the inventive structures. For example,values for the feedback and input resistors R_(f) and R₁ could bechanged to obtain higher or lower gains. Also, an active resistivesummer could be constructed to sum only the measured signal V_(tx) andthe replica signal V_(txr) (or a high-pass version of the replica), forexample. Additionally, while the communication channel has beendescribed with respect to a twisted pair of cables, the invention mayalso be practiced with other communication channels such as optical andwireless channels. Moreover, this invention should not be limited togigabit transmission rates and can be practiced at any transmission raterequiring the signal processing characteristics of the invention. Ofcourse, these and other such modifications are covered by the presentinvention.

1. A transmit canceller in a communication channel, the channelincluding a first transceiver for transmitting and receiving signals anda replica transmitter for generating an input replica transmissionsignal, a composite signal at a near end comprising a transmissionsignal of the first transceiver and a signal received from a secondtransceiver, said transmit canceller comprising: an operationalamplifier having a first polarity input terminal, a second polarityinput terminal, and an output terminal; a feedback element incommunication with the second polarity input terminal and the outputterminal; a first input resistor in communication with the secondpolarity input terminal and the measured signal input; a second inputresistor in communication with the second polarity input terminal andthe replica signal input; and a predetermined voltage source incommunication with the first polarity input terminal of the operationalamplifier, wherein the received signal is an output at the outputterminal of the operational amplifier.
 2. The transmit cancelleraccording to claim 1, further comprising a connection between a baselinecorrection current source and the second polarity input terminal.
 3. Thetransmit canceller according to claim 2, further comprising a chargepump to control the baseline correction current source.
 4. The transmitcanceller according to claim 3, wherein the charge pump controls thebaseline correction current source based on an error between anequalized baseline signal and a sliced baseline signal.
 5. The transmitcanceller according to claim 2, further comprising a connection betweena common-mode shift current source and the second polarity inputterminal.
 6. The transmit canceller according to claim 1, wherein thereplica signal comprises a high pass signal.
 7. The transmit cancelleraccording to claim 1, wherein the replica signal comprises a negativesignal and the transmit canceller further includes a third resistor incommunication with the second polarity input terminal and a low passpositive replica signal input.
 8. The transmit canceller according toclaim 1, wherein the communication channel comprises a gigabit channel.9. A transmit canceller in a communication channel, the channelincluding a first transceiver for transmitting and receiving signals anda replica transmitter for generating an input replica transmissionsignal, a composite signal at a near end comprising a transmissionsignal of the first transceiver and a signal received from a secondtransceiver, said transmit canceller comprising: means for amplifyingincluding a first polarity input terminal, a second polarity inputterminal, and an output terminal; feedback means for communicating withthe second polarity input terminal and the output terminal; first meansfor communicating with the second polarity input terminal and themeasured signal input; second means for communicating with the secondpolarity input terminal and the replica signal input; and means forsupplying a predetermined voltage to the first polarity input terminalof the amplifying means, wherein the received signal is an output at theoutput terminal of the amplifying means.
 10. The transmit cancelleraccording to claim 9, further comprising means for correcting baselinewander.
 11. The transmit canceller according to claim 10, furthercomprising means for pumping to control the baseline correcting means.12. The transmit canceller according to claim 11, wherein the means forpumping controls the baseline correction means based on an error betweenan equalized baseline signal and a sliced baseline signal.
 13. Thetransmit canceller according to claim 10, further comprising means forconnecting a common-mode shift current source and the second polarityinput terminal.
 14. The transmit canceller according to claim 9, whereinthe replica signal comprises a high pass signal.
 15. The transmitcanceller according to claim 9, wherein the replica signal comprises anegative signal and the transmit canceller further includes a thirdmeans for communicating with the second polarity input terminal and alow pass positive replica signal input.
 16. The transmit cancelleraccording to claim 9, wherein the communication channel comprises agigabit channel.
 17. A method of reducing a transmission signal from acomposite signal in a communication channel, the channel including afirst transceiver for transmitting and receiving signals and a replicatransmitter for generating an input replica transmission signal, thecomposite signal at a near end comprising a transmission signal of thefirst transceiver and a signal received from a second transceiver, saidmethod comprising the steps of: providing an operational amplifierhaving a first polarity input terminal, a second polarity inputterminal, and an output terminal; arranging a feedback element to be incommunication with the second polarity input terminal and the outputterminal; arranging a first resistive element to be in communicationwith the second polarity input terminal and the measured signal input;arranging a second resistive element to be in communication with the esecond polarity input terminal and the replica signal input; arranging apredetermined voltage source to be in communication with the firstpolarity input terminal of the operational amplifier; and outputting asignal at the output terminal that reduces the transmission signal. 18.The method according to claim 17, further comprising a step ofconnecting a baseline correction current source to the second polarityinput terminal of the operational amplifier.
 19. The method according toclaim 18, further comprising a step of controlling the baselinecorrection current source with a charge pump.
 20. The method accordingto claim 19, wherein the charge pump controls the baseline correctioncurrent source based on an error between an equalized baseline signaland a sliced baseline signal.
 21. The method according to claim 18,further comprising a step of connecting a common-mode shift currentsource to the second polarity input terminal to control a common-modevoltage of the operational amplifier.
 22. The method according to claim17, wherein the replica signal comprises a high pass signal.
 23. Themethod according to claim 17, wherein the replica signal comprises anegative signal and the transmit canceller further includes a thirdresistive element in communication with the second polarity inputterminal and a low pass positive replica signal input.
 24. The methodaccording to claim 17, wherein the communication channel comprises agigabit channel.
 25. An electrical circuit for reducing a transmissionsignal comprising: an active resistive summer having an operationalamplifier that includes a first polarity input terminal, a secondpolarity input terminal, and an output terminal, said active resistivesummer further comprising: a feedback element in communication with theoutput terminal and the second polarity input terminal; a first resistorin communication with the second polarity input terminal and a compositesignal, the composite signal having a transit signal component and areceive signal component; and a second resistor in communication withthe second polarity input terminal and a replica of the transmit signal.